DETAILED ACTION
Notice of Pre-AIA or AIA Status
The present application, filed on or after March 16, 2013, is being examined under the first inventor to file provisions of the AIA .
Response to Amendment
The amendment to the claims filed on 05/22/2026 complies with the requirements of 37 CFR 1.121(c) and has been entered. Claims 1-2, 7-9, and 12-14 are amended.
Response to Arguments
Applicant's Arguments/Remarks filed 05/22/2026 (hereinafter Resp.) have been fully considered hereinafter.
Applicant’s main argument is that Taher et al., U.S. Patent Application Publication No. 2018/0159637 (hereinafter Taher1) does not teach the amended limitation “by shifting the time sample window relative to the reference signal, ... in order to achieve a common time sample distribution for each channel of the multi-channel RF receiving system” whereby “Support for the amendments can be seen from Figures 3 and 9 of the as filed specification” – See Resp.,p.2:¶2, because “[i]nstead of shifting the sampling window itself to correct this, Taherl applies a post-processing Jilter (a Time Domain Equalizer) to the already-captured data to digitally estimate time alignment, Taher1 at para. [0073]” – See id.,¶3. Examiner respectfully disagrees with Applicant’s understanding of Figures 3 and 9 and further of the time equalization method of Taher1 which is no different from the method 100 disclosed in the present Specification and Figure 1.
First, taking as example Figure 3 of the present disclosure, “[i]n detail, Figure 3 shows an actual sample distribution 31 around a reference point 32 in a pre-determined sample area” that “may further be compared with a desired sample distribution so to determine a difference between the desired and the actual sample distribution 31 so compare difference between the channels” concluding that “the sample for the channel of Figure 3 does not need any adjusting” because “the samples 31 are evenly distributed around the reference point 32” – See Spec., p.9:7-15. On the contrary, “by adjusting the channels in Figure 4 in accordance with the method 100, the two channels may be synchronized” – See Spec., p.10:2-3, whereby “Figure 1 illustrates a flowchart of a method for 100 for synchronizing time alignment . . . the method comprises the step of determining 103 propagation time difference and digital synchronization error between each channel” and “the method adjusts 105 the channels in the time domain in accordance with the determined adjustment parameters of synchronization for each channel” – See Spec., p8:3-15 (emphasis added). Therefore, there is no “shifting the time sample window” disclosed in Figure 3 and/or the method 100 when applied. Similarly, “Figure 9 shows the channels prior to calibration/synchronization in accordance with the present disclosure,” i.e., method 100 in Figure 1, whereby “the digital video data channel 1 and channel 2 before delay calibration . . . has a differing signal in the time sample windows” shown in block A, upper level, and “in order to adjust the delay/errors in accordance with the present disclosure, the propagation time difference and digital synchronization error may be determined 103 (or e.g. in accordance with the embodiment comprising the step of 103a-103d) so to derive adjustment parameters,” i.e., method 100 is applied – See id. at 5-15, wherein “after determining propagation error and digital synchronization error/difference between the channels (e.g. by method step 103 or 103a-103d), the channels may be adjusted so to be synchronized i.e. resulting in that the sample windows are centered around a reference point (evident from reference letters B' and B")” – See id. at 19-23. To be sure, the Specification is clear about “the method utilizing means to detect position of the reference signal within a time sample window”– See Spec., p.3:23-24, not vice versa, i.e., shifting the time window to be centered around the reference signal, as argued. Therefore, the sample windows being centered around the reference signal is a result of applying the propagation error and digital synchronization error/difference to the samples by practicing the method 100, i.e., there is no shifting the time sample window relative to the reference signal that would produce the results of method 100.
Second, the detailed description of Figures 3 and Figure 9, together with the method 100 disclosed in Figure 1, map directly to the disclosures in Taher1. For example, Figure 3 maps precisely over Figure 6, showing 3 channels that are time-aligned on the reference signal whereby the sample window is centered on the peak value corresponding to each channel’s response to the reference signal center frequency, i.e., the time duration of the sample window (e.g., a FFT window) is limited to the center frequency and most important frequency ripples. Figure 9 maps to Figure 5 showing that the 3 channels are not time-aligned, i.e., before applying the time and frequency equalization methods. Furthermore, Taher1 discloses that “Frequency domain equalization and correlation typically require timing synchronization to determine the placement of the FFT window” – See [¶0058], i.e., to center the time-sampling window around the peak (correlation) value in Fig. 6. Therefore, Taher1 teaches “timing synchronization is based on either autocorrelation of the received signal or its cross-correlation with the original sequence” and this synchronization is performed prior to frequency-equalization – See id. Thus, Applicant’s argument supra that “[i]nstead of shifting the sampling window itself to correct this, Taherl applies a post-processing Jilter (a Time Domain Equalizer) to the already-captured data to digitally estimate time alignment, Taher1 at para. [0073]” is first inapposite to the very method disclosed in the present Specification in accord with Taher1 because none of them is shifting the sampling window itself to correct the time offsets of different receiving channels; and second, misinterprets the very method disclosed when the Specification clearly states that “The step of determining propagation time difference and digital synchronization error may comprises for each channel, determining an actual sample distribution around a reference point in a pre-determined sample area,” – See Spec., p.5:7-9 whereby “The detector unit may generate a detector signal proportional to a time delay with the formula . . . over N reference signal samples” – See id., at 23-25 when a person of ordinary skills in the art would appreciate that determining a distribution of samples happens “after the fixed sampling windows have closed” using “mathematical filtering” – cf. Resp., p.2:¶3 (arguing that “While Taherl may eventually produce aligned data outputs, it does so entirely through mathematical filtering after the fixed sampling windows have closed. Taherl never actually shifts the time sample window relative to the reference signal to achieve a common sample distribution as explicitly required by amended Claim 1. By physically shifting the time sample window, the subject matter of amended Claim 1 enables capturing of the correct data at the source, rather than attempting to mathematically correct misaligned data post-capture” when there is no “physically shifting the time sample window” disclosed in the present Application but rather the same procedure disclosed in greater detail in Taher11)
While the Office recognizes that the Applicant has a duty to zealously advocate for the client, the arguments used should be rooted in the present disclosure as understood by a person of ordinary skills in the art. Here, no reasonable person of ordinary skills in the art would understand from the disclosure how to physically shift a time sample window while maintaining the reference signal, as argued. Signals are transitory in time, therefore the reference signal start point is captured through correlation with known sequences as disclosed in Taher1 (again, in details not covered by the present Specification) and not by casting a butterfly net (e.g. a filter) that moves to catch the reference signal at the desired point in time.
In sum, Applicant’s arguments are not persuasive.
Claim Objections
Amended Claims 1 and 12 are objected to because of the following informalities: the meaning of "a common time sample distribution," as illustrated in Figures 3 and 9, is unclear to a person of ordinary skills in the art; “the same distribution of time samples” as shown in Fig. 3 or “the same time sample” as shown in Fig. 9 represent a better description of the intended meaning. Appropriate correction is required because the Specification does not define “a common time sample distribution” as claimed, and Figures. 3 and 9 each represents a different aspect of the claimed invention having in common only the result of synchronization using the adjusted parameters obtained by processing the reference signal.
Claim Rejections - 35 USC § 112 (a)
The following is a quotation of the first paragraph of 35 U.S.C. 112(a):
(a) IN GENERAL.—The specification shall contain a written description of the invention, and of the manner and process of making and using it, in such full, clear, concise, and exact terms as to enable any person skilled in the art to which it pertains, or with which it is most nearly connected, to make and use the same, and shall set forth the best mode contemplated by the inventor or joint inventor of carrying out the invention.
Amended Claims 1 and 12 and their dependents are rejected under 35 U.S.C. 112(a) first paragraph, as failing to comply with the written description requirement. The claim(s) contains subject matter which was not described in the specification in such a way as to reasonably convey to one skilled in the relevant art that the inventor or a joint inventor, at the time the application was filed, had possession of the claimed invention.
The Amended independent Claims now require to “adjust the channels in the time domain by shifting the time sample window relative to the reference signal.” Applicant’s Arguments/Remarks Resp. supra emphasizes the requirement means “physically shifting the time sample window” – See Resp., p.2:¶3. However, neither the Specification nor the drawings disclose how to physically shift a time window to be centered on a reference signal. On the contrary, the Specification states that “after determining propagation error and digital synchronization error/difference between the channels (e.g. by method step 103 or 103a-103d), the channels may be adjusted so to be synchronized i.e. resulting in that the sample windows are centered around a reference point (evident from reference letters B' and B")” – See Spec., p.8:19-23. To be sure, the Specification is clear about “the method utilizing means to detect position of the reference signal within a time sample window”– See Spec., p.3:23-24, not vice versa, i.e., physically shifting the time window to be centered around the reference signal, as argued.
For these reasons, Amended Claims 1 and 12 and their dependent claims are rejected under 35 U.S.C. §112(a) for lack of sufficient description and new matter added by amendment.
.
Claim Rejections - 35 USC § 102
The following is a quotation of the appropriate paragraphs of 35 U.S.C. 102 that form the basis for the rejections under this section made in this Office action:
A person shall be entitled to a patent unless –
(a)(2) the claimed invention was described in a patent issued under section 151, or in an application for patent published or deemed published under section 122(b), in which the patent or application, as the case may be, names another inventor and was effectively filed before the effective filing date of the claimed invention.
Claim(s) 1-2, 5, and 7-13, as amended, are rejected under 35 U.S.C. 102(a)(2) as being anticipated by Taher et al., U.S. Patent Application Publication No. 2018/0159637 (hereinafter Taher1).
Regarding Claim 1, Taher1 teaches a method for synchronizing time alignment in a multi-channel radio frequency (RF) receiving system (“systems, storage media, and methods for calibrating a multiple input multiple output (MIMO) radio system” – See [¶0004], including “a method whereby a single calibration signal is used by a MIMO radio system to perform each of time synchronization, phase synchronization, and frequency response correction for each of the multiple receivers” – See [¶0005] and Figs. 1 and 10 whereby “a wide-band complex pilot signal is routed to the multiple N receivers” that “may digitize the received pilot signal” – See [¶0006] and each receiver may “derive an equalizer that determines the time offset, phase offset, and frequency response of the channel associated with the receiver” – See [¶0037]; in addition, a “10 MHz common reference signal 108 ensures that the digital to-analog converter (DAC) of the signal generator 120 and the receivers' digitizer analog-to-digital converters (ADCs) 102-106 are locked with respect to the reference and to each other” so “the time alignment in the digitized ADC samples will hold for a long duration after applying the timing alignment calibration method described” – See [¶0033]), the method comprising
injecting an amplitude modulated or phase modulated reference signal into each channel in the multi-channel RF receiving system at a location associated with each antenna input (“a pilot waveform signal may be generated at the Cal signal generator 120” – See [¶0066] and Fig. 2 wherein the pilot is injected at a location associated with each antenna input Antenna 1-3, whereby a “radio frequency (RF) up-conversion is then performed to prepare the signal for transmission. The signal is then put through a 1:N splitter (1:3 splitter in the particular embodiment shown in FIG. 3), and one of the split signals is transmitted to one of the receivers 102-106” – See [¶0067] and the “complex pilot signal that is wideband, has desirable autocorrelation properties, and has a spectrum that is mostly flat throughout its bandwidth” e.g., “cyclically shifted versions of a ZC sequence imposed on a signal result in zero correlation with each other at the receiver because cyclically shifted versions of a ZC sequence are orthogonal,” i.e., phase modulated – See [¶0035]; furthermore the reference signal may experience “frequency distortion as the frequency response of each receive hardware may be different. Additionally, there may be a phase offset between the complex samples received in each receiver” – See [¶0036]),
detecting a position of the reference signal within a time sample window for each channel after analog to digital conversion (“correlation typically require timing synchronization to determine the placement of the FFT window,” e.g., “based on either autocorrelation of the received signal or its cross-correlation with the original sequence” – See [¶0058] and Figs. 4- 5 showing the Time Domain Equalizer wherein, e.g., in Fig. 4 at 312 “a processor at the receiver finds the Pilot Start, which comprises a coarse timing of the signal” after analog to digital conversion – See [¶0069] “so that the equalizer training can start from that sample” – See [¶0075] i.e., the time sample window start point, and “FIG. 5 shows channel impulse response data (in dBm) for impulses obtained through correlation for 3 channels, without employing the calibration techniques” wherein “The peaks in each of the 3 impulse response charts are a few samples wide and they do not perfectly overlap with each other” – See [¶0080], i.e., the end of the time sample windows ends where CIR Amplitude samples are below, e.g., -110dB)
determining propagation time difference and digital synchronization error between each channel within electronics of the multi-channel RF receiving system (“FIG. 4 is illustrated for a single channel, but the steps illustrated in FIG. 4 may be repeated for each channel in the MIMO radio system” – See [¶0064] wherein, after determining the “coarse timing of the [reference] signal,” i.e., the coarse propagation time difference as shown in Fig. 5; then, “at 314, a Reference Pilot (i.e., the known local copy of the calibration signal) is used to adaptively equalize until convergence is obtained, outputting equalizer coefficients” – See [¶0069], i.e., the digital synchronization error which are applied in Operation Mode shown in Fig. 4, wherein “these equalizer coefficients may be used by a filter DSP block that operates after the ADC. At the output of the time domain equalizer, calibrated digital samples of the ADC acquired waveform may be produced that are time aligned down to picoseconds” – See [¶0073]);
determining adjustment parameters, for synchronizing time alignment, for each channel (as shown in Fig. 4, in Operation Mode, “[o]nce the signal is received by one of the radios 102-106, the radio again performs RF down-conversion on the signal 322, analog-to-digital conversion 324” and “The coarse timing recover block may not be required” – See [¶0072] but, for each channel, “at 328, the signal may be filtered with the equalizer coefficients derived at 314 in operation mode” – See [¶0073]); and
adjusting the channels in the time domain by shifting the time sample
to the reference signal in accordance with the determined adjustment parameters of
synchronization for each channel in order to achieve a common time sample distribution2 for
each channel of the multi-channel RF receiving system (first, “FIG. 5 shows channel impulse response data (in dBm) for impulses obtained through correlation for 3 channels, without employing the calibration techniques” while “FIG. 6 shows channel impulse response data (in
dBm) for impulses obtained through correlation for 3 channels, after calibration has been performed according to one embodiment described” wherein “the correlation peak is narrow and one sample wide” for all channels, therefore by shifting a 1 sample time window at the 3 channels peaks in Fig. 5 with the calculated difference and projecting that difference to several samples around the peak, the aligned–i.e., common time at peak CIR value–Channel Impulse Response in time domain of Fig. 6 is obtained; furthermore, because “the time domain equalizer may be advantageous when the inter-symbol interference (ISI) caused by the non-flat frequency response of the imperfect receiver hardware is several data symbols long” – See [¶0073], i.e., high frequency ripples are significative as in Fig. 7, then frequency domain equalization may be necessary, and that comes after time equalization method because “Frequency domain equalization and correlation typically require timing synchronization to determine the placement of the FFT window,” i.e., “prior timing synchronization [] based on either autocorrelation of the received signal or its cross-correlation with the original sequence” as described supra – See [¶0058]; “[t]he equalizer may be configured to operate on the fast Fourier transform (FFT) of the digitized samples, i.e., it may operate in the frequency domain” on the output of the Time Domain Equalizer to further remove non-flat frequency response of the imperfect receiver due to “frequency distortion as the frequency response of each receive hardware may be different” and “phase offset between the complex samples received in each receiver” – See [¶0036]; here, in Calibration Mode as shown in Fig. 3, “The equalizer may be configured to operate on the fast Fourier transform (FFT) of the digitized samples, i.e., it may operate in the frequency domain” – See [¶0045] in a FFT window over which “during Operation Mode, an FS-FDE digital signal processing (DSP) block is inserted at 258” and “an inverse fast Fourier transform (IFFT) is performed on the processed signal at 260” to obtain time-aligned and frequency-equalized samples on each channel– See [¶0063]).
Therefore, Amended Claim 1 is anticipated by Taher1.
Regarding Claim 2, dependent from Amended Claim 1, Taher1 further teaches the method according to claim 1, wherein the channels are adjusted in the time domain by at least one of:
a coarse delay shift in the order of an integer number of analog-to-digital conversion-(ADC), samples (“a processor at the receiver finds the Pilot Start, which comprises a coarse timing of the signal” – See [¶0068] and Fig. 5, showing that “peaks in each of the 3 impulse response charts are a few samples wide and they do not perfectly overlap with each other,” – See [¶0080] but “after calibration has been performed . . . the channels are now time aligned” because “the correlation peak is narrow and one sample wide,” – See [¶0081]; furthermore, the Time of Arrival (TOA) corresponding to the Pilot Start is around 10 samples delay as shown in Fig. 5 or 8 samples as in Fig. 6, determined using “cross correlation of the received ZC/CAZAC sequence with a local copy of the sequence [and] may begin at any point (without prior timing synchronization), e.g., because of the periodic nature of the transmitted signal)” and “allow determination of the TOA” – See [¶0058]); and
a fine delay shift in the order of a fractional of ADC samples in accordance with the determined parameters for synchronization for each channel (the coarse delay shift “is used to adaptively equalize until convergence is obtained, outputting equalizer coefficients” – See [¶0069] whereby convergence is obtained with samples that are “time aligned down to picoseconds” – See [¶0073] i.e., fractional of an ADC sample, as shown in FIG.6 wherein samples are aligned at delay shifts of far less than 1ns).
Therefore, Claim 2 is anticipated by Taher1.
Regarding Claim 5, dependent from Amended Claim 1, Taher1 further teaches the method according to claim 1, wherein adjusting the channels comprise using an interpolation filter (“since the equalizer is half spaced, the time-domain samples may be routed to a decimating de-interleaver at 218 that outputs two streams of time domain samples, each operating at half rate” used to calculate equalizer coefficients – See [¶0045] and Fig. 3 showing also the decimating de-interleaver 248 before performing equalization on the received RF signals, i.e., “the decimating de-interleaver 248 may route the incoming time domain samples alternately to the even and odd sample paths 250-252” – See [¶0062] followed by a digital adjustment comprising the Fractionally Spaced-Frequency Domain Equalizer using “the same coefficients that were derived during the Cal Mode at 230” and “calibrated digital samples of the ADC acquired waveform may be produced that are time aligned down to picoseconds” – See [¶0063]; a person of ordinary skills in the art would appreciate that a “decimating de-interleaver” is a down-sampling filter within the meaning of an interpolation filter in the present application3).
Therefore, Claim 5 is anticipated by Taher1.
Regarding Amended Claim 7, dependent from Amended Claim 1, Taher1 further teaches the method according to claim 1, wherein adjusting the channels comprise at least one of using a shift register, and controlling a clock generation circuit of an analog-to-digital conversion (ADC) to adjust the phase of an outgoing signal (“With the 10 MHz clocks locked, the time alignment in the digitized ADC samples will hold for a long duration after applying the timing alignment calibration method” – See [¶0033] whereby “[a] further enhancement to this method may involve daisy chaining the Local Oscillators (LOs) that do the down conversion to ensure that the phase alignments also hold for a long duration after the disclosed method for calibration is complete” – See [¶0034]).
Therefore, Amended Claim 7 is anticipated by Taher1.
Regarding Amended Claim 8, dependent from Amended Claim 1, Taher1 further teaches the method according to claim 1, wherein the synchronization is performed at startup of the multi-channel RF receiving system (“after completion of the calibration process, the MIMO system may be configured to transition to an Operation Mode. After transitioning from the Cal Mode to Operation Mode, . . . the switches 110 may connect the radios 102-106 to the antennas 112-116, to allow the radios to receive signals from the antennas,” – See [¶0070] i.e., the synchronization is performed before putting the multi-channel RF receiving system in operation, e.g., at startup because although “an acquisition ADC start trigger may be shared between each receiver during Cal Mode and during Operation mode . . . there is likely to be several nanoseconds of residual timing misalignment between one ADC and another-hence the equalizer derived during Cal Mode may remove this timing mismatch and also phase align the receivers. After switching to the Operation Mode, the timing and phase alignment functions of the calibration equalizer previously derived may hold if all the N receivers' ADCs are initiated for acquisition together via the shared trigger” – See [¶0074]).
Therefore, Amended Claim 8 is anticipated by Taher1.
Regarding Amended Claim 9, dependent from Amended Claim 1, Taher1 further teaches the method according to claim 1, wherein the synchronization is checked at pre-set intervals during operation of the multi-channel RF receiving system (“the data communication protocol used by the radios may be designed to automatically repeat the calibration step at pre-set intervals to improve MIMO performance,” e.g., “there may be time slots for calibration where data packets are not sent, but where real-time calibration of the phase, frequency and timing alignment is performed” – See [¶0076]).
Therefore, Amended Claim 9 is anticipated by Taher1.
Regarding Claim 10, dependent from Amended Claim 1, Taher1 further teaches the method according to claim 1, wherein the step of determining propagation time difference and digital synchronization error comprises:
for each channel, determining an actual sample distribution around a reference point in a pre-determined sample area in each of said time sample windows (when “[a] wideband complex correlation signal is received simultaneously by the multiple receivers through the switches … [e]ach receiver runs a correlator and the resulting correlation peaks or channel impulse responses have been plotted as a function of offset” around peak value(s) – See [¶0078], whereby “without employing the calibration techniques . . . [t]here are multiple peaks above the horizontal dotted line caused by the different frequency responses in each receiver” and “[t]he peaks in each of the 3 impulse response charts are a few samples wide and they do not perfectly overlap with each other” in a time sample window of around 0.2µs wherein the reference point is the maximum peak value – See [¶0080] and Fig. 5);
setting a desired sample distribution having evenly distributed samples around said reference point (e.g., distribution of “channel impulse response data (in dBm) for impulses obtained through correlation for 3 channels, after calibration has been performed . . . all the plots align very closely at the peak, showing excellent correlation” and “the correlation peak is narrow and one sample wide” with evenly distributed samples around the peak/reference point – See [¶0081] and Fig. 6);
determining a difference between each desired sample distribution and each actual sample distribution for each channel (“Reference Pilot (i.e., the known local copy of the calibration signal) is used to adaptively equalize until convergence is obtained, outputting equalizer coefficients” – See [¶0069] and step 314 in Fig. 4, wherein a “the time-domain equalization filter” determines the difference between each desired sample distribution of the reference pilot signal, sampled with the 10MHz clock, and each actual sample distribution for each channel obtained from the ADC 310 with the same sample period in the time domain sample window of around 0.2µ– See [¶0073] and Fig. 4);
comparing the difference between the channels (“The equalizer coefficients . . . remove the residual nanosecond timing offset between the various channels” – See [¶0057], i.e., the time-domain equalization filter above compares the differences between channels, e.g., to remove a common denominator or to avoid “a long convergence time during calibration” if one difference is too large – See [¶0073]).
Therefore, Claim 10 is anticipated by Taher1.
Regarding Claim 11, dependent from Claim 10, further teaches the method according to claim 10, wherein the reference point is a center point in said pre-determined sample area (e.g., as shown in Fig. 6, “the correlation peak is narrow and one sample wide” and “all the plots align very closely at the peak” – See [¶0081], i.e., the reference point is a center point in the pre-determined sample area of one periodicity, e.g., 100ns, of the sampling/reference signal).
Therefore, Claim 11 is anticipated by Taher1.
Regarding Amended Claim 12, Taher 1 teaches in Fig. 1 a multi-channel radio-frequency (RF) receiving system (Rx 102-106) for synchronizing time alignment in different receiver channels, the RF receiving system comprising: a plurality of antennas having antenna inputs (Antenna 1-3); a plurality of receiver channels (Switches 110); control circuitry (when implemented in a computer system, the receiving system is “configured to include a processor (or a set of processors)” executing program instructions “to implement any of the various method embodiments described” – See [¶0110]); wherein the control circuitry is configured to: perform the steps of Claim 1, received with the same language. Because Claim 1 is anticipated by Taher1, Claim 12 is also anticipated by Taher1.
Regarding Amended Claim 13, teaches a non-transitory computer-readable storage medium storing one or more programs configured to be executed by one or more control circuitry of a multi-channel RF receiving system (“the present invention may be realized as a computer-implemented method, a computer-readable memory medium, or a computer system” – See [¶0108] whereby “non-transitory computer readable memory medium may be configured so that it stores program instructions and/or data, where the program instructions, if executed by a computer system, cause the computer system to perform a method, e.g., any of a method embodiments described herein, or, any combination of the method embodiments described herein. or. any subset of any of the method embodiments described herein, or, any combination of such subsets” – See [¶0109]) the one or more programs comprising instructions for performing the method according to claim 1. Because Claim 1, as amended, is anticipated by Taher1, Amended Claim 13 is also anticipated by Taher1.
Therefore Claims 1-2, 5, and 7-13, as amended, are rejected under 35 U.S.C. §102(a)(2) as anticipated by Taher1.
Claim Rejections - 35 USC § 103
The following is a quotation of 35 U.S.C. 103 which forms the basis for all obviousness rejections set forth in this Office action:
A patent for a claimed invention may not be obtained, notwithstanding that the claimed invention is not identically disclosed as set forth in section 102, if the differences between the claimed invention and the prior art are such that the claimed invention as a whole would have been obvious before the effective filing date of the claimed invention to a person having ordinary skill in the art to which the claimed invention pertains. Patentability shall not be negated by the manner in which the invention was made.
The factual inquiries for establishing a background for determining obviousness under 35 U.S.C. 103 are summarized as follows:
1. Determining the scope and contents of the prior art.
2. Ascertaining the differences between the prior art and the claims at issue.
3. Resolving the level of ordinary skill in the pertinent art.
4. Considering objective evidence present in the application indicating obviousness or nonobviousness.
This application currently names joint inventors. In considering patentability of the claims the examiner presumes that the subject matter of the various claims was commonly owned as of the effective filing date of the claimed invention(s) absent any evidence to the contrary. Applicant is advised of the obligation under 37 CFR 1.56 to point out the inventor and effective filing dates of each claim that was not commonly owned as of the effective filing date of the later invention in order for the examiner to consider the applicability of 35 U.S.C. 102(b)(2)(C) for any potential 35 U.S.C. 102(a)(2) prior art against the later invention.
Claims 3-4 are rejected under 35 U.S.C. 103 as being unpatentable over Taher1 as applied to Amended Claim 1 above, and further in view of Mirhaj et al., U.S Patent Publication No. 10312927 (hereinafter Mirhaj).
Regarding Claim 3, dependent from Claim 1, Taher1 further teaches the method according to claim 1, wherein the reference signal is periodic signal (e.g., in determining the time window, “cross correlation of the received [reference signal] with a local copy of the sequence may begin at any point (without prior timing synchronization), e.g., because of the periodic nature of the transmitted signal, which may simplify signal processing architecture” – See [¶0058]). Although Taher1 teach the reference signal may be of other forms (“other wideband complex sequences may be used in other embodiments instead of or in addition to ZC sequences” – See [¶0035], Thaler1 does not teach a saw tooth reference signal.
Mirhaj, like Taher1, teaches multi-channels calibration such as “time-interleaved ADCs [that] suffer from mismatch between different channels of the ADC” – See col.3:35-36, of which “timing skew may be especially difficult to detect and correct” – See id.:62-63, and further “provide[s] techniques and apparatus for calibrating time-interleaved ADCs using a unique set of signals, which may be generated on chip ( e.g., on the same integrated circuit as the ADC itself)” – See id.:66-67, col.4:1-2. Mirhaj teaches “The signal generator of FIG. 2 includes a pattern generator 202 and a conversion circuit 204” – See col.5:40-42 whereby “[t]he conversion circuit 204 (also referred to as a signal generating circuit or "sig_gen") is configured to receive the waveform bitstream 220 from the pattern generator 202 and convert the bitstream to an analog signal for inputting into the time-interleaved ADC circuit 100 at the ADC input node” for calibration – See col.6:16-23.
Mirhaj further teaches this reference signal is a saw tooth signal (“A suitable waveform bitstream may include a relatively higher frequency component (e.g., to aid timing skew calibration) combined with a lower frequency component ( e.g., to aid with gain calibration). For example, the waveforms may include a double sawtooth signal (two ramps), a single sawtooth signal (a single ramp)” – See col.6:42-47 and Fig. 4, wherein “the double sawtooth signal 401 (or more specifically, one period of the double sawtooth signal 401) includes a low-frequency portion (e.g., a negative slope line corresponding to the ramp-down portion 402 and a positive slope line corresponding to the ramp-up portion 404), combined with a high-frequency portion (e.g., the jagged deviations from the negative and positive slope lines, 5 due to the bit patterns of {0,1,0} and {1,0,1}). By having both low and high frequency components, the double sawtooth signal 401 may be suitable for both gain and timing skew calibrations of the time-interleaved ADC” – See col.6:66-67, col.7:1-9)
Thus, Taher1 and Mirhaj each teaches techniques and apparatus for multi-channel calibration to remove timing skewness between channels, i.e., for synchronizing time alignment among the channels using a reference signal injected into an ADC for calibration purposes. A person of ordinary skill in the art before the effective filing date of the claimed invention would have understood that the saw-tooth like calibration waveform taught in Mihraj could have been used with the Time Domain Equalizer and method taught in Taher1 because both use a pilot waveform signal at the input of a multichannel ADC sampled then with a high frequency common clock to determine a propagation time difference between channels. Furthermore, a person of ordinary skill in the art would have been able to carry out the combination through techniques known in the art. Finally, the combination achieves the predictable result of obtaining very precise digital synchronization errors/coefficients as taught in Taher1 while being able to calibrate for amplitude differences at the same time, as taught in Mirhaj.
Thus, Claim 3 is obvious over Taher1 in view of Mirhaj.
Regarding Claim 4, dependent from Claim 3, Taher1 in view of Mirhaj further teaches the method according to claim 3, wherein the saw tooth signal has a rise and fall time in the order of 1μs (“the waveform bitstream generating circuit 210 may be configured to generate a 127-bit bitstream corresponding to a double sawtooth signal” – See Mirhaj:col.6:50-52 and Fig. 4 whereby the ADC clock may have 10 MHz as taught in Taher1:[¶0033](“ The 10 MHz common reference signal 108 ensures that the digital-to- analog converter (DAC) of the signal generator 120 and the receivers' digitizer analog-to-digital converters (ADCs) 102-106 are locked with respect to the reference and to each other. With the 10 MHz clocks locked, the time alignment in the digitized ADC samples will hold for a long duration after applying the timing alignment calibration method described”), i.e., the 127 bits are decoded with a clock of 10MHz that yields a rise and fall time of a double saw-tooth signal of about 6.3μs, i.e., in the order of 1μs4).
Therefore, Claim 4 is obvious over Taher1 in view of Mirhaj.
In sum, Claims 3-4 are rejected under 35 U.S.C. §103 as obvious over Taher1 in view of Mirhaj.
Claim 6 is rejected under 35 U.S.C. 103 as being unpatentable over Taher1 as applied to claim5 above, and further in view of Gudovskiy et al., U.S. Patent Application No. 2018/0316482 (hereinafter Gudovskiy).
Regarding Claim 6, dependent from Claim 5, although Taher1 teaches the method according to claim 5, wherein the interpolation filter operates down sampling in an interpolation module, Taher1 does not teach that the interpolation filter operates also the steps of up-sampling and sample delay in the interpolation module.
Gudovskiy, like Taher1, teaches “[s]ystems and methods . . . in which a wireless receiver can be configured to digitally synchronize a receive sampling rate to a transmit sampling rate,” and further teaches “a digital interpolator controlled by a timing control unit with a timing offset estimator,” whereby, like block 314 in Taher1 that derives the finer granularity equalizer coefficients, “[t]he timing control unit can be configured to calculate and output parameters to the digital interpolator” – See [¶0023]. Furthermore, the digital interpolator in Gudovskiy is a programmable delay block or a customizable equalization filter for sub-samples delays because “[t]he timing offset estimator can be configured to calculate and provide to the timing control unit a sampling period ratio control word and an instantaneous timing offset control word” – See [¶0023]. Therefore, Gudovskiy teaches the interpolation filter operates also the steps of up-sampling, down-sampling and sample delay in the interpolation module because the digital interpolation filter uses the sampling period ratio control word to adapt upwards or downwards the sampling frequency of the receiver and the instantaneous timing offset control word to adapt its sub-sample delay (“interpolation filter 442 can be implemented as a "modified" version of a Farrow filter. The Farrow structure can be used realize an adjustable, fractional-delay, FIR (finite-length impulse response) filter” – See [¶0059] and Fig. 10)
Thus, Taher1 and Gudovskiy each teaches systems and methods for correcting for a timing offset in a sampling signal of the receiver relative to a reference sampling signal using digital adjustments based on equalizer coefficients or control words applied to a digital filter buffering received signal samples. A person of ordinary skill in the art before the effective filing date of the claimed invention would have understood that the digital interpolator taught by Gudovskiy could have been substituted in for the programable filter controlled with equalizer coefficients on each of the multiple input multiple output (MIMO) radio channels taught by Taher1 because each of these blocks provides for reducing time offsets between the multiple channels using measured or estimates from a time synchronization control unit. Furthermore, a person of ordinary skill in the art would have been able to carry out the substitution through techniques known in the art. Finally, the substitution achieves the predictable result of applying cheaper digital interpolation methods that allow calibration of MIMO time synchronization through sampling rate correction in the digital domain, as taught by Gudovskiy.
Therefore, Claim 6 is rejected under 35 U.S.C. §103 as obvious over Taher1 in view of Gudovskiy.
Amended Claim 14 is rejected under 35 U.S.C. 103 as being unpatentable over Taher1 as applied to Amended Claim 12 above, and further in view of Papazian et al., "A Radio Channel Sounder for Mobile Millimeter-Wave Communications: System Implementation and Measurement Assessment," in IEEE Transactions on Microwave Theory and Techniques, vol. 64, no. 9, pp. 2924-2932, Sept. 2016, doi: 10.1109/TMTT.2016.2592530 (hereinafter Papazian).
Regarding Amended Claim 14, dependent from Amended Claim 12, Taher1 further teaches a vehicle comprising the RF receiving system according to claim 12 (“the MIMO radio device may be a measurement apparatus designed to perform channel sounding or other wireless measurements using cellular or another wireless technology,” e.g., mounted on a vehicle “for measuring radio channel conditions for cellular MIMO communications (e.g., 5G or NR communications)” – See [¶0025]).
Although it is general knowledge that vehicles have continuous wireless connectivity providing synchronization through GSSN/GPS and perform channel sounding5 to correct for differences in parameter estimates such as delay spread, delay window, or correlation bandwidth, Taher1 does not specifically disclose a vehicle comprising the RF receiving system.
Papazian discloses “the first sounder that is capable of mobile measurements at mm-wave frequencies” with a “delay resolution of the system is 1 ns” whereby “[t]he RX array is mounted on a location-aware robot, which is battery operated” – See Abstract, p2924, and “satisfy scenarios envisioned for 5G networks, such as massive-MIMO antenna arrays, vehicular speeds in device-to-device communications [10], and transmission of high-bandwidth signals” – See §I, col1:¶3, p 2925. Like Taher1, Papazian’s receiver uses “[d]igital correlation-based processing [18], [19] in which a high-speed digitizer acquires the received intermediate-frequency (IF) signal and correlation is performed in postprocessing. This reduces the measurement time of the channel impulse response to the duration of the codeword” – See id. Papazian further uses “[a] 16-element receive-antenna array oriented in both the azimuth and elevation directions whose field-of-view covers the upper hemisphere . . . allowing measurement of the impulse response across all 16 elements within 65.5μs” that “provides a maximum coherence time corresponding to vehicular speeds up to approximately 100 km/h” – See id., and “a portable rubidium clock for timing synchronization, with negligible drift over the 65.5-μs rotation. The receive system is global-positioning-system (GPS)-equipped for outdoor operation and robotically navigated with a laser range finder for indoor applications” – See §I, col2:¶1, p 2925 and Fig. 1 (c), p 2926.
Thus, Taher1 and Papazian each discloses a MIMO RF receiver equipped for measuring/calibrating time synchronization delays with sub-nanosecond precision. A person of ordinary skill in the art before the effective filing date of the claimed invention would have understood that the improvements to the MIMO RF receiver for channel sounding measurement/calibration when mounted on a vehicle, as taught in Papazian, could have been applied to the MIMO RF receiver of Taher1 because they share the same schematic based on digital correlation-based processing of a reference signal, including detection of the start block/codeword in the signal. Furthermore, a person of ordinary skill in the art would have been able to carry out the improvement through techniques known in the art. Finally, the improvement achieves the predictable result of expanding the application of the common schematic to vehicular mobility using 5G network, as taught by Papazian.
Therefore, Amended Claim 14 is rejected under 35 U.S.C. §103 as obvious over Taher1 in view of Papazian.
Conclusion
The prior art made of record and not relied upon is considered pertinent to applicant's disclosure:
Taher et al., U.S. patent Application Publication No. 2019/0149198 as referenced in previous Office Action;
PETTERSSON et al, EPO Patent Application Publication No: EP 1293043 B1, discloses digital calibration of ADCs in which the calibration may be accomplished with dynamic estimation of reference signals that have unknown parameters and/or that have amplitude swings that exceed the full swing of the ADC;
Teng., China Patent Application Publication No.: CN112737591A discloses system and method for multichannel signal synchronization and comprises an adjustable digital delay module, a digital-to-analog converter and an adjustable analog delay module which are sequentially arranged on all channels respectively;
Yang, China Patent Application Publication No.: CN109361389A discloses a multi-channel mismatch error calibration method and system for a time interleaved analog to digital converter;
Carney et al., U.S. Patent Publication No. 5,537,435 discloses multichannel wireless communication transceiver architecture employing overlap and add or polyphase signal processing functionality, for wideband signal processing, including sample time adjustment;
Bierly et al., U.S. Patent Publication No. 6,421,372 discloses a parallel digital matched filter which performs numerous simultaneous correlations of a received spread spectrum signal against various replica offsets of its spreading sequence;
Fernandez et al., U.S. Patent Publication No. 7,148,828 discloses a method for calibrating time interleaved samplers comprising applying a calibration signal to a time-interleaved sampling device;
Pipon et al., Singapore Patent Application Publication No. SG1020161094 discloses methods of estimating interference noise using learning sequences known by the receiver, e.g., to carry out anti-jamming equalization using frequency interpolation;
Zheng et al, China Patent Application Publication No. CN104297738 discloses synchronization calibration device and synchronization calibration and error compensation method for multi-channel receiver;
Rossi, "Influence of measurement conditions on the evaluation of some radio channel parameters," in IEEE Transactions on Vehicular Technology, vol. 48, no. 4, pp. 1304-1316, July 1999, doi: 10.1109/25.775378;
Harris et al., "Multirate digital filters for symbol timing synchronization in software defined radios," in IEEE Journal on Selected Areas in Communications, vol. 19, no. 12, pp. 2346-2357, Dec. 2001, doi: 10.1109/49.974601;
Ferreira et al., Real-time high-resolution radio frequency channel sounder based on the sliding correlation principle. IET Microwave. Antennas Propagation, 9: 837-846, 2015, https://doi.org/10.1049/iet-map.2014.0165;
Seijo et al., "Portable Full Channel Sounder for Industrial Wireless Applications With Mobility by Using Sub-Nanosecond Wireless Time Synchronization," in IEEE Access, vol. 8, pp. 175576-175588, 2020, doi: 10.1109/ACCESS.2020.3025896.
Applicant's amendment necessitated the new ground(s) of rejection presented in this Office action. Accordingly, THIS ACTION IS MADE FINAL. See MPEP § 706.07(a). Applicant is reminded of the extension of time policy as set forth in 37 CFR 1.136(a).
A shortened statutory period for reply to this final action is set to expire THREE MONTHS from the mailing date of this action. In the event a first reply is filed within TWO MONTHS of the mailing date of this final action and the advisory action is not mailed until after the end of the THREE-MONTH shortened statutory period, then the shortened statutory period will expire on the date the advisory action is mailed, and any nonprovisional extension fee (37 CFR 1.17(a)) pursuant to 37 CFR 1.136(a) will be calculated from the mailing date of the advisory action. In no event, however, will the statutory period for reply expire later than SIX MONTHS from the mailing date of this final action.
Any inquiry concerning this communication or earlier communications from the examiner should be directed to LUCIA GHEORGHE GRADINARIU whose telephone number is (571)272-1377. The examiner can normally be reached Monday-Friday 9:00am - 5:00pm EST.
Examiner interviews are available via telephone, in-person, and video conferencing using a USPTO supplied web-based collaboration tool. To schedule an interview, applicant is encouraged to use the USPTO Automated Interview Request (AIR) at http://www.uspto.gov/interviewpractice.
If attempts to reach the examiner by telephone are unsuccessful, the examiner’s supervisor, Joseph AVELLINO can be reached at (571)272-3905. The fax phone number for the organization where this application or proceeding is assigned is 571-273-8300.
Information regarding the status of published or unpublished applications may be obtained from Patent Center. Unpublished application information in Patent Center is available to registered users. To file and manage patent submissions in Patent Center, visit: https://patentcenter.uspto.gov. Visit https://www.uspto.gov/patents/apply/patent-center for more information about Patent Center and https://www.uspto.gov/patents/docx for information about filing in DOCX format. For additional questions, contact the Electronic Business Center (EBC) at 866-217-9197 (toll-free). If you would like assistance from a USPTO Customer Service Representative, call 800-786-9199 (IN USA OR CANADA) or 571-272-1000.
/L.G.G./ Examiner, Art Unit 2478
/JOSEPH E AVELLINO/ Supervisory Patent Examiner, Art Unit 2478
1 To be sure, Taher1 discloses how the corrections coefficients are used – See [¶0073] (“these equalizer coefficients may be used by a filter DSP block that operates after the ADC”). A person of ordinary skills in the art would know that a “filter DSP” whose role is to compensate for time-domain offset is a time-shifting filter applied to the samples of the signal. This is no different to the method disclosed by the present Application providing “adjusting the channels in the time domain in accordance with the determined adjustment parameters of synchronization for each channel” – See Spec., p.3:9-10, whereby adjusting the channels in the time domain is reasonably understood by one of ordinary skills in the art as applying a time-shifting filter. Moreover, this filtering is a mathematical operation on digital samples using “digital synchronization error” calculated “after analog to digital conversion of said reference signal” – See id.:2-4 (emphasis added), therefore the present Application is also “attempting to mathematically correct misaligned data post-capture.”
2 “Common time sample distribution” is interpreted as a time-aligned distribution of digital samples as shown in Fig. 3 of the present disclosure, i.e, the time samples are overlapping as much as possible in time as shown in Fig. 6 of Taher1 when compared with Fig. 5 of Taher1.
3 See, e.g., Specification, 4:23-24 (stating “interpolation filter may operate steps of up-sampling, sample delay, and down-sampling”). This is different from the general knowledge in the art whereby interpolation is the operation of up-sampling and decimation is the operation of down-sampling and whereby filters capable of performing both operations and also fractional delay on input signals are called polyphase filters – See, e.g., Section III, Harris et al., "Multirate digital filters for symbol timing synchronization in software defined radios," in IEEE Journal on Selected Areas in Communications, vol. 19, no. 12, pp. 2346-2357, Dec. 2001, doi: 10.1109/49.974601.
4 Mirhaj teaches the method to determine the number of bits of the reference signal waveform as “[t]he waveform bitstream 22 may have a bit pattern with a total number of bits that shares no common factor with the number of channels 106 (e.g., m=8)” – See col.5:45-48, i.e., a calculation is available to have a saw-tooth signal with precisely 1 μs slopes.
5 See, e.g., Seijo et al., "Portable Full Channel Sounder for Industrial Wireless Applications With Mobility by Using Sub-Nanosecond Wireless Time Synchronization," in IEEE Access, vol. 8, pp. 175576-175588, 2020, doi: 10.1109/ACCESS.2020.3025896
Prosecution Insights