DETAILED ACTION
Notice of Pre-AIA or AIA Status
1. The present application, filed on or after March 16, 2013, is being examined under the first inventor to file provisions of the AIA .
In the response to this office action, the Examiner respectfully requests that support be shown for language added to any original claims on amendment and any new claims. That is, indicate support for newly added claim language by specifically pointing to page(s) and line numbers in the specification and/or drawing figure(s). This will assist the Examiner in prosecuting this application.
Information Disclosure Statement
2. The information disclosure statement filed on December 26, 2024 and February 09, 2026 have been considered and placed in the application file.
Drawings
3. The drawings are objected to because: Reference label “10E” in figure 6 missing; microphones “30” and “30-1” in figure 9 having speaker symbols.
Corrected drawing sheets in compliance with 37 CFR 1.121(d) are required in reply to the Office action to avoid abandonment of the application. Any amended replacement drawing sheet should include all of the figures appearing on the immediate prior version of the sheet, even if only one figure is being amended. The figure or figure number of an amended drawing should not be labeled as “amended.” If a drawing figure is to be canceled, the appropriate figure must be removed from the replacement sheet, and where necessary, the remaining figures must be renumbered and appropriate changes made to the brief description of the several views of the drawings for consistency. Additional replacement sheets may be necessary to show the renumbering of the remaining figures. The replacement sheet(s) should be labeled “Replacement Sheet” in the page header (as per 37 CFR 1.84(c)) so as not to obstruct any portion of the drawing figures. If the changes are not accepted by the examiner, the applicant will be notified and informed of any required corrective action in the next Office action. The objection to the drawings will not be held in abeyance.
Claim Rejections - 35 USC § 102
4. The following is a quotation of the appropriate paragraphs of 35 U.S.C. 102 that form the basis for the rejections under this section made in this Office action:
A person shall be entitled to a patent unless –
(a)(1) the claimed invention was patented, described in a printed publication, or in public use, on sale, or otherwise available to the public before the effective filing date of the claimed invention.
5. Claim 1 is rejected under 35 U.S.C. 102(a)(1) as being anticipated by Kates U.S. Patent Application Publication 20060008101.
Regarding claim 1, Kates teaches an electronic device (Hearing aid 100 is comprised of an input transducer 101, preferably a microphone, an analog-to-digital (A/D) converter 103, a digital processing means 105 (e.g., a digital signal processor or DSP), a digital-to-analog (D/A) converter 107, and an output transducer 109, preferably a speaker, Fig. 1, par [0026], see Kates) comprising:
a microphone configured to receive a first acoustic sound (i.e., audio signals) and generate a first input signal based on the first acoustic sound (In operation, input transducer 101 receives audio signals and converts the audio signals into analog electrical signals, Fig. 1, par [0026], see Kates);
filter circuitry (A block diagram of a frequency-warped processing system according to the invention is shown in FIG. 7, the processing system being applicable to the digital processing means of an audio system such as the hearing aid shown in FIG. 1, see Fig. 7, par [0034], see Kates) comprising a plurality of filter stages, wherein each filter stage of the plurality of filter stages comprises an all-pass filter (In at least one embodiment of the invention, the processing system is a warped FIR side-branch system consisting of a 16-sample data buffer and a 32-point warped FFT used in conjunction with a 31-tap warped FIR filter. The input data segment, after passing through the all-pass stages that provide the frequency warping, is windowed with a 32-point hanning (von Hann) window and no frequency-domain smoothing is applied to the spectrum, par [0040], see Kates) and a multiplier (see symbol “X” in Fig. 7, also see “Multiply input” in last step of Fig. 8, see Kates), and wherein the filter circuitry is configured to:
determine a filter input based on the first input signal (The incoming signal x(n) (701) is passed through a cascade of first-order all-pass filters 703 of the form given by Eq. (1), see Fig. 7, par [0034], see Kates); and
process the filter input via the plurality of filter stages to generate a filtered signal (The system output is then calculated by convolving the delayed samples with the compression gain filter (711): y .function. ( n ) = k = 0 K .times. g k .function. ( n ) .times. p k .function. ( n ) ( Eq . .times. 12 ) ##EQU10## where {g.sub.k(n)} are the compression filter coefficients, see equation (12), par [0035], see Kates); and
a speaker configured to output a second acoustic sound based on the filtered signal (The analog electrical signals are converted by A/D converter 103 into digital electrical signals that are subsequently processed by DSP 105 to form a digital output signal. The digital output signal is converted by D/A converter 107 into an analog electrical signal. The analog signal is used by output transducer 109, e.g., a speaker, to produce an audio signal that is heard by the user of hearing aid 100, see Fig. 7, par [0034], see Kates).
Kates thus teaches all the claimed limitations.
Claim Rejections - 35 USC § 103
6. In the event the determination of the status of the application as subject to AIA 35 U.S.C. 102 and 103 (or as subject to pre-AIA 35 U.S.C. 102 and 103) is incorrect, any correction of the statutory basis for the rejection will not be considered a new ground of rejection if the prior art relied upon, and the rationale supporting the rejection, would be the same under either status.
7. The following is a quotation of 35 U.S.C. 103 which forms the basis for all obviousness rejections set forth in this Office action:
A patent for a claimed invention may not be obtained, notwithstanding that the claimed invention is not identically disclosed as set forth in section 102 of this title, if the differences between the claimed invention and the prior art are such that the claimed invention as a whole would have been obvious before the effective filing date of the claimed invention to a person having ordinary skill in the art to which the claimed invention pertains. Patentability shall not be negated by the manner in which the invention was made.
8. This application currently names joint inventors. In considering patentability of the claims the examiner presumes that the subject matter of the various claims was commonly owned as of the effective filing date of the claimed invention(s) absent any evidence to the contrary. Applicant is advised of the obligation under 37 CFR 1.56 to point out the inventor and effective filing dates of each claim that was not commonly owned as of the effective filing date of the later invention in order for the examiner to consider the applicability of 35 U.S.C. 102(b)(2)(C) for any potential 35 U.S.C. 102(a)(2) prior art against the later invention.
9. Claims 2, and 9-10 are rejected under 35 U.S.C. 103 as being unpatentable over Kates U.S. Patent Application Publication 20060008101 in view of Oliver U.S. Patent Application Publication 20100266143.
Regarding claim 2, Kates teaches the electronic device of claim 1. Kates teaches wherein the plurality of filter stages is configured to process the filter input based on operations (In at least one embodiment of the invention, the processing system is a warped FIR side-branch system consisting of a 16-sample data buffer and a 32-point warped FFT used in conjunction with a 31-tap warped FIR filter. The input data segment, after passing through the all-pass stages that provide the frequency warping, is windowed with a 32-point hanning (von Hann) window and no frequency-domain smoothing is applied to the spectrum, par [0040], see Kates) .
However, Kates does not explicitly disclose comprising: multiplying, via the multiplier of a first filter stage of the plurality of filter stages, the filter input by a respective coefficient of the multiplier of the first filter stage to generate a first transitional signal; adding the first transitional signal to a first all-pass output of the all-pass filter of a second filter stage of the plurality of filter stages to generate a second transitional signal; and applying the all-pass filter of the first filter stage to the second transitional signal to generate a second all-pass output signal.
Oliver teaches frequency-warped audio equalizer (see Title) in which comprising: multiplying, via the multiplier of a first filter stage of the plurality of filter stages, the filter input by a respective coefficient of the multiplier of the first filter stage (Four multipliers 640, 642, 644, and 646 are shown in communication with the all-pass filters 601-611. These multipliers represent coefficients h.sub.0, h.sub.1, h.sub.2, and h.sub.3 of the equalization filter 600b, Fig. 6B, par [0070], see Oliver) to generate a first transitional signal (Each filter 601-611 in the example equalization filter 600b includes the same warping factor .rho., as represented by multipliers 448, 504, 514, 604, 612, and 620 (corresponds to a first transitional signal), Fig. 6B, par [0073], see Oliver); adding the first transitional signal to a first all-pass output of the all-pass filter of a second filter stage of the plurality of filter stages to generate a second transitional signal (n more detail, the output 637 of the filter 607 is combined with the output 633 of the filter 603 by adder 630, which provides an output to the multiplier 642 (coefficient h.sub.1) (corresponds to a second transitional signal), Fig. 6B, par [0071], see Oliver); and applying the all-pass filter of the first filter stage to the second transitional signal to generate a second all-pass output signal (Likewise, the output 639 of the filter 609 is combined with the output 631 of the filter 601 and is provided to adder 632, which provides an output to the multiplier 644 (coefficient h.sub.2), Fig. 6B, par [0071], see Oliver). Since “the outputs of the filters 601-611 are therefore superimposed together (see Fig. 6B, par [0071])” thus output of each adder 630, 632 being considered as “transitional signal”.
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the frequency-warped audio equalizer taught by Oliver with the electronic device of Kates such that to obtain comprising: multiplying, via the multiplier of a first filter stage of the plurality of filter stages, the filter input by a respective coefficient of the multiplier of the first filter stage to generate a first transitional signal; adding the first transitional signal to a first all-pass output of the all-pass filter of a second filter stage of the plurality of filter stages to generate a second transitional signal; and applying the all-pass filter of the first filter stage to the second transitional signal to generate a second all-pass output signal in order to accomplish the same task with improved accuracy on the same processor, as suggested by Oliver in paragraph [0086].
Regarding claim 9, Kates teaches the electronic device of claim 1. However, Kates does not explicitly disclose which wherein the filter circuitry is configured to process the filter input in fixed point.
Oliver teaches frequency-warped audio equalizer (see Title) in which advantageously, in certain embodiments the equalization filter 800 may be implemented in a fixed-point processor. In addition, divisions by 2 can be replaced by shifts. Thus, the equalization filter 800 can be robust numerically, even for low-resolution fixed-point math operations, par [0089], see Oliver).
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the frequency-warped audio equalizer taught by Oliver with the electronic device of Kates such that to obtain wherein the filter circuitry is configured to process the filter input in fixed point in order to accomplish the same task with improved accuracy on the same processor, as suggested by Oliver in paragraph [0086].
Regarding claim 10, Kates teaches the electronic device of claim 1. However, Kates does not explicitly disclose wherein the filter circuitry is configured to perform a multiplication for the multiplier of each filter stage of the plurality of filter stages via a shift-and-add operation.
Oliver teaches frequency-warped audio equalizer (see Title) in which. multiplication operations can be eliminated in one embodiment by reducing the number of digits of precision used for the warping factors and/or coefficients. By using a few bits of precision, for example, multiplications can be replaced with additions and shifts. In addition, divisions by 2 can be replaced by shifts. Thus, the equalization filter 800 can be robust numerically, even for low-resolution fixed-point math operations par [0089], see Oliver).
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the frequency-warped audio equalizer taught by Oliver with the electronic device of Kates such that to obtain wherein the filter circuitry is configured to perform a multiplication for the multiplier of each filter stage of the plurality of filter stages via a shift-and-add operation in order to accomplish the same task with improved accuracy on the same processor, as suggested by Oliver in paragraph [0086].
10. Claim 3 is rejected under 35 U.S.C. 103 as being unpatentable over Kates U.S. Patent Application Publication 20060008101 in view of Jensen et al. U.S. Patent Application Publication 20140086425 (hereinafter, “Jensen”).
Regarding claim 3, Kates teaches the electronic device of claim 1. However, Kates does not explicitly disclose comprising an error microphone configured to receive a mixed acoustic sound comprising the first acoustic sound and the second acoustic sound.
Jensen teaches active noise cancellation using multiple reference microphone signals (see Title) in which comprising an error microphone (error microphone 3, Fig. 2, par [0027], see Jensen) configured to receive a mixed acoustic sound comprising the first acoustic sound and the second acoustic sound (The adaptive algorithm engine or controller 9 may implement a gradient decent algorithm, e.g. least mean squares (LMS), which is designed to find the proper state or digital filter coefficients that tends to minimize the residual noise or error between the anti-noise sound and the ambient or background noise. This error is reflected in a signal that is derived from the output of an error microphone 3, Fig. 2, par [0027], see Jensen); wherein each controller is to adjust a respective one of the adaptive filters using an adaptive algorithm engine (e.g., a gradient descent algorithm engine such as a least means squares, LMS, algorithm), based on input from the respective reference microphone signal and an error signal derived from an error microphone output (par [0006], see Jensen)).
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the active noise cancellation using multiple reference microphone signals taught by Jensen with the electronic device of Kates such that to obtain comprising an error microphone configured to receive a mixed acoustic sound comprising the first acoustic sound and the second acoustic sound in order to help improve the user's listening experience by attempting to produce a quieter environment, as suggested by Jensen in paragraph [0003].
Kates in view of Jensen, as modified, teaches wherein the filter circuitry is configured to adjust a respective coefficient of the multiplier of one or more filter stages of the plurality of filter stages based on the mixed acoustic sound, adjust a respective all-pass coefficient of the all-pass filter of the filter stage based on the mixed acoustic sound, or both (The state of each adaptive filter 4a, 4a including its digital filter coefficients is repeatedly updated by a respective adaptive filter algorithm engine of an adaptive filter controller 9a, 9b (Fig. 2, par [0027], see Jensen); coefficients of the adaptive filters are adjusted based on input from the reference signals and based on input from a signal from an error microphone (par [0047], see Jensen). The all-pass filter used for frequency warping is given by: A( z ) = z - 1 - a 1 - a .times. .times. z - 1 ( Eq . .times. 1 ), see (Eq. 1) where a is the warping parameter. For a real, setting a>0 increases the frequency resolution at low frequencies and setting a<0 increases the resolution at high frequencies (par [0027], see Kates)). The motivation is in order to help improve the user's listening experience by attempting to produce a quieter environment, as suggested by Jensen in paragraph [0003].
11. Claim 4, and 6 are rejected under 35 U.S.C. 103 as being unpatentable over Kates U.S. Patent Application Publication 20060008101 in view of Massie et al. U.S. Patent 8611551 (hereinafter, “Massie”).
Regarding claim 4, Kates teaches the electronic device of claim 1. Kates further teaches wherein the filter circuitry is configured to generate the filter input (The incoming signal x(n) (701) is passed through a cascade of first-order all-pass filters 703 of the form given by Eq. (1), see Fig. 7, par [0034], see Kates).
However, Kates does not explicitly disclose the filter input based on: converting the first input signal from a first sampling rate to a second sampling rate less than the first sampling rate to generate an intermediate signal; and modulating the intermediate signal to generate the filter input, wherein the filter input comprises a third sampling rate less than the first sampling rate.
Massie teaches low latency active noise cancellation system (see Title) in which the filter input based on: converting the first input signal from a first sampling rate to a second sampling rate less than the first sampling rate to generate an intermediate signal (The digital reference signal R(n) is provided to decimator module 408, hereinafter also referred to as a decimator. The decimator module 408 downsamples the digital reference signal R(n) to produce a decimated digital reference signal R'(n) at a second sampling rate less than the first sampling rate. In other words, the decimator module 408 downsamples the digital reference signal R(n) by a predetermined downsampling factor (decimation factor) to form the decimated digital reference signal R'(n), Fig. 4, col. 7, lines 40-48, see Massie); and modulating the intermediate signal to generate the filter input (In operation, the analog reference signal r(t) generated by the reference microphone 106 is provided to oversampling data converter 406 within the A/D module 400. The oversampling data converter 406 converts the analog reference signal r(t) into a digital reference signal R(n) at a first sampling rate (Fig. 4, col. 7, lines 22-26, see Massie)), wherein the filter input comprises a third sampling rate less than the first sampling rate (This approach is different from other embodiments described above, in which a decimator is used to downsample the stream generated by the oversampling data converter. This downsampling may use several sample rate conversion stages and yield multi-bit data (e.g., 8, 16, 24 bits) as described above (col. 10, line 63 – col 11, line 1 see Massie)).
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the low latency active noise cancellation system taught by Massie with the electronic device of Kates such that to obtain the filter input based on: converting the first input signal from a first sampling rate to a second sampling rate less than the first sampling rate to generate an intermediate signal; and modulating the intermediate signal to generate the filter input, wherein the filter input comprises a third sampling rate less than the first sampling rate in order to provide more flexibility and functionality than traditional analog systems, as suggested by Massie in col. 11, lines 10-12.
Regarding claim 6, Kates teaches the electronic device of claim 1. However, Kates does not explicitly disclose wherein the first acoustic sound is at a first location of the microphone at a first time and has traveled to a second location of the speaker at a second time, wherein the microphone is configured to receive the first acoustic sound at the first time, and wherein the speaker is configured to output the second acoustic sound at the second time.
Massie teaches low latency active noise cancellation system (see Title) in which in a typical feedforward ANC system, a reference microphone provides a reference signal based on the background noise captured at a reference position (corresponds to a first time) (col. 1, lines 40-43, see Massie); The analog circuitry filters and inverts the analog reference signal received from the reference microphone to form an analog compensation signal, which is then provided to the loudspeaker (corresponds to a second time) (col. 2, lines 4-8, see Massie). An analog electric signal g(t), which is formed by converting the digital noise reduction signal F'(n), and optionally a digital desired signal S(n) from the audio device 104, is then provided to the audio transducer 116 (col. 5, lines 57-60, see Massie).
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the low latency active noise cancellation system taught by Massie with the electronic device of Kates such that to obtain wherein the first acoustic sound is at a first location of the microphone at a first time and has traveled to a second location of the speaker at a second time, wherein the microphone is configured to receive the first acoustic sound at the first time, and wherein the speaker is configured to output the second acoustic sound at the second time in order to provide more flexibility and functionality than traditional analog systems, as suggested by Massie in col. 11, lines 10-12.
12. Claims 5 is rejected under 35 U.S.C. 103 as being unpatentable over Kates U.S. Patent Application Publication 20060008101 in view of Massie et al. U.S. Patent 8611551 (hereinafter, “Massie”), and further in view of Kerner et al. U.S. Patent Application Publication 20150023415 (hereinafter, “Kerner”).
Regarding claim 5, Kates in view of Massie teaches the electronic device of claim 4. Kates in view of Massie, as modified, teaches wherein the filter circuitry comprises a delta sigma modulator or a sigma delta modulator (As described below, the analog reference signal r(t) is converted into a decimated digital reference signal R'(n) using an oversampling data converter such as a sigma-delta modulator, col. 5, lines 42-45, see Massie).
However, Kates in view of Massie does not explicitly disclose wherein the delta sigma modulator or the sigma delta modulator is configured to modulate the intermediate signal with an n-bit quantizer greater than one, and wherein the filter input comprises an n-bit depth.
Kerner teaches method for noise shaping and a noise shaping filter (see Title) in which both ADCs and DACs can employ delta-sigma modulation. A delta-sigma ADC first encodes an analog signal using delta-sigma modulation and then applies a digital filter to form a higher-resolution digital output (par [0020], see Kerner). In the following, quantization stages, quantization errors and bit-depth reduction are described. Quantization is the process of mapping a large set of input values to a smaller set such as rounding values to some unit of precision, e.g. by reducing a bit-depth of the input values. A device or a stage that performs quantization is called a quantizer (par [0018], see Kerner). The quantizing operation may be a bit-depth reduction or a rounding operation as described above, e.g. a bit-depth reduction by 3 bits according to Q(x)=round(x/8). The error e(n-1) 108 is obtained by left-shifting the output y(n) obtaining a shifted output with a bit-depth of 15 bits and subtracting the delayed internal value v(n-1) from that shifted output (see Fig. 1, par [0033], see Kerner).
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the method for noise shaping and a noise shaping filter taught by Kerner with the electronic device of Kates in view of Massie such that to obtain wherein the delta sigma modulator or the sigma delta modulator is configured to modulate the intermediate signal with an n-bit quantizer greater than one, and wherein the filter input comprises an n-bit depth in order to improve the conversion, as suggested by Kerner in paragraph [0020].
13. Claims 7-8 are rejected under 35 U.S.C. 103 as being unpatentable over Kates U.S. Patent Application Publication 20060008101 in view of Park et al. U.S. Patent Application Publication 20170229110 (hereinafter, “Park”).
Regarding claim 7, Kates teaches the electronic device of claim 1. However, Kates does not explicitly disclose wherein the second acoustic sound is based on a mixed audio signal comprising at least a portion of the filtered signal and at least a portion of another audio signal.
Park teaches systems, methods, apparatus, and computer-readable media for adaptive active noise cancellation (see Title) in which the ANC apparatus will typically also include a digital-to-analog converter (DAC) arranged to convert antinoise signal SY10 to analog form upstream of loudspeaker LS10. As noted below, it may also be desirable for the ANC apparatus to mix a desired sound signal with the antinoise signal (in either the analog or digital domain) to produce an audio output signal for reproduction by loudspeaker LS10. Examples of such desired sound signals include a received (i.e. far-end) voice communications signal, a music or other multimedia signal, and a sidetone signal (Figs. 1A, 1B, par [0076], see Park).
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the systems, methods, apparatus, and computer-readable media for adaptive active noise cancellation taught by Park with the electronic device of Kates such that to obtain wherein the second acoustic sound is based on a mixed audio signal comprising at least a portion of the filtered signal and at least a portion of another audio signal for purpose of enhancing or separating clear desired sound from background sounds originating from multiple directions, as suggested by Park in paragraph [0182].
Regarding claim 8, Kates in view of Park teaches the electronic device of claim 7. Kates in view of Park, as modified, teaches wherein the other audio signal comprises a pre-recorded audio track, a generated audio track distinct from the first input signal, or a combination thereof (In one such example, desired sound signal SD10 is a reproduced audio signal, such as a far-end voice communications signal (e.g., a telephone call) or a multimedia signal (e.g., a music signal, which may be received via broadcast or decoded from a stored file), par [0104], see Park). The motivation is for purpose of enhancing or separating clear desired sound from background sounds originating from multiple directions, as suggested by Park in paragraph [0182].
14. Claims 11, and 17-19 are rejected under 35 U.S.C. 103 as being unpatentable over Massie et al. U.S. Patent 8611551 (hereinafter, “Massie”) in view of Kates U.S. Patent Application Publication 20060008101, and further in view of Davenport et al. "The Pros and Cons of Compressive Sensing for Wideband Signal Acquisition: Noise Folding versus Dynamic Range", IEEE Transactions on Signal Processing, Vol. 60, No. 9, September 2012, (pp. 4628-4642), 15 pages (hereinafter, “Davenport”).
Regarding claim 11, Massie teaches filter circuitry (As a result, low latency active noise cancellation is performed utilizing digital filter circuitry which is not subject to the inaccuracies and drift of analog filter components. In doing so, the present technology provides robust, flexible, and high quality active noise cancellation, col. 2, lines 33-37, see Massie) configured to:
receive an input signal at a first sampling rate and a first bit-depth (In step 1020, the analog reference signal r(t) is converted into the digital reference signal R(n) using an oversampling data converter. The oversampling data converter produces a single-bit (i.e., bit-depth is one-bit) data stream, which is fed directly into a filter, Fig. 10, col. 13, lines 55-58, see Massie);
convert the first sampling rate and the first bit-depth of the input signal to a second sampling rate less than the first sampling rate and a second bit-depth (This approach is different from other embodiments described above, in which a decimator is used to downsample the stream generated by the oversampling data converter. This downsampling may use several sample rate conversion stages and yield multi-bit data (e.g., 8, 16, 24 bits) (i.e., bit-depth) as described above. This multi-bit data used in other approaches should be distinguished from the single-bit data stream described herein (Fig. 4, col. 10, line 63 – col. 11 line 3, see Massie) to generate an intermediate signal (see output of decimator 408, Fig. 4 inside low latency ANC processing system 210, see Massie);
modulate the intermediate signal to generate a filter input signal comprising a processing rate less than the first sampling rate (The digital reference signal R(n) is provided to decimator module 408, hereinafter also referred to as a decimator. The decimator module 408 downsamples the digital reference signal R(n) to produce a decimated digital reference signal R'(n) at a second sampling rate less than the first sampling rate. In other words, the decimator module 408 downsamples the digital reference signal R(n) by a predetermined downsampling factor (decimation factor) to form the decimated digital reference signal R'(n) (Fig. 4, col. 7, lines 40-48, see Massie)).
However, Massie does not explicitly disclose filter, via a warped finite impulse response filter, the filter input signal based on a set of filter coefficients to generate a filtered signal; and output the filtered signal.
Kates teaches spectral enhancement using digital frequency warping (see Title) in which As the gain versus frequency function is a set of pure real numbers, the inverse FFT of the warped time-domain filter results in a set of filter coefficients that is real and has even symmetry (709). The time-domain filter coefficients are then windowed to reduce ripple in the compressor output when the system is excited by a swept sinusoid (Fig. 7, par [0035], see Kates).
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the spectral enhancement using digital frequency warping taught by Kates with the filter circuitry of Massie such that to obtain filter, via a warped finite impulse response filter, the filter input signal based on a set of filter coefficients to generate a filtered signal; and output the filtered signal for purpose of improving ease of listening and possibly speech intelligibility, as suggested by Kates in Abstract.
However, Massie in view of Kates does not explicitly disclose a second bit-depth greater than the first bit-depth.
Davenport teaches The Pros and Cons of Compressive Sensing for Wideband Signal Acquisition: Noise Folding versus Dynamic Range in which this experiment highlights the very real benefit of reduced sampling rates; easing the sampling rate requirement can allow us to use higher fidelity hardware components, such as high bit-depth quantizers (page 4638, right column, next-to-last paragraph, see Davenport).
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the “The Pros and Cons of Compressive Sensing for Wideband Signal Acquisition: Noise Folding versus Dynamic Range” taught by Davenport with the filter circuitry of Massie in view of Kates such that to obtain a second bit-depth greater than the first bit-depth in order to significant improvements in the dynamic range of the system, as suggested by Davenport in page 4634, left column, second paragraph.
Regarding claim 17, this claim merely specifies a non-transitory, machine-readable medium comprising instructions, wherein, when executed by one or more processors, the instructions cause the one or more processors to control operations of filter circuitry or to perform the operations of claim 11 and is therefore interpreted and rejected under Massie in view of Kates in view of Davenport for the same reasons. It is noted that Massie in view of Kates in view of Davenport teaches a non-transitory computer readable storage medium as described herein has embodied thereon a program executable by a processor to perform a method for reducing an acoustic energy level at a listening position as described above (see col. 13, lines 6-9, see Massie).
Regarding claim 18, Massie in view of Kates in view of Davenport teaches the non-transitory, machine-readable medium of claim 17. Massie in view of Kates in view of Davenport, as modified, teaches wherein the input signal comprises a first bit-depth and the filter input signal comprises a second bit-depth greater than the first bit-depth (This approach is different from other embodiments described above, in which a decimator is used to downsample the stream generated by the oversampling data converter. This downsampling may use several sample rate conversion stages and yield multi-bit data (e.g., 8, 16, 24 bits) as described above. This multi-bit data used in other approaches should be distinguished from the single-bit data stream described herein (Fig. 4, col. 10, line 63 – col. 11 line 3, see Massie). This experiment highlights the very real benefit of reduced sampling rates; easing the sampling rate requirement can allow us to use higher fidelity hardware components, such as high bit-depth quantizers (page 4638, right column, next-to-last paragraph, see Davenport). The motivation is in order to significant improvements in the dynamic range of the system, as suggested by Davenport in page 4634, left column, second paragraph.
Regarding claim 19, Massie in view of Kates in view of Davenport teaches the non-transitory, machine-readable medium of claim 17. However, Massie in view of Kates in view of Davenport, ad modified, teaches wherein modulating the intermediate signal comprises noise shape quantizing the intermediate signal via an n-bit quantizer greater than one (Fig. 1. A midrise uniform quantization function Qb(x-i) with bits, saturation level G, and quantization interval Δ, see Fig. 1, page 3634, right column, see Davenport).
15. Claims 12-14 are rejected under 35 U.S.C. 103 as being unpatentable over Massie et al. U.S. Patent 8611551 (hereinafter, “Massie”) in view of Kates U.S. Patent Application Publication 20060008101 in view of Davenport et al. “The Pros and Cons of Compressive Sensing for Wideband Signal Acquisition: Noise Folding versus Dynamic Range”, IEEE Transactions on Signal Processing, Vol. 60, No. 9, September 2012, 4628-4642, 15 pages (hereinafter, “Davenport”), and further in view of Cabler U.S. Patent 5414424.
Regarding claim 12, Massie in view of Kates in view of Davenport teaches the electronic device of claim 11. Massie in view of Kates in view of Davenport, as modified, teaches wherein the filter circuitry comprises a delta sigma modulator or a sigma delta modulator (As described below, the analog reference signal r(t) is converted into a decimated digital reference signal R’(n) using an oversampling data converter such as a sigma-delta modulator, col. 5, lines 42-45, see Massie); the delta sigma modulator or the sigma delta modulator configured to modulate the intermediate signal (Digital sigma-delta module 620 then quantizes (i.e., reduces the number of bits) the digital output of the CIC interpolator 610 and shapes the quantization noise (col. 9, lines 55-57, see Massie)) with a quantizer greater than one bit to noise shape the intermediate signal (this experiment highlights the very real benefit of reduced sampling rates; easing the sampling rate requirement can allow us to use higher fidelity hardware components, such as high bit-depth quantizers (page 4638, right column, next-to-last paragraph, see Davenport).
However, Massie in view of Kates in view of Davenport does not explicitly disclose wherein quantization noise is shifted away from a first frequency range and to a second frequency range higher than the first frequency range.
Cabler teaches fourth-order cascaded sigma-delta modulator (see Title) in which the use of higher order sigma-delta modulators has become desirable in many applications for several reasons. One reason is because the introduction of higher order modulators increases the number of integrations to be carried out, which results in a decrease in the noise level of the passband as the quantization noise is shifted to a higher frequency level. Another reason is because the use of higher order modulators keeps the oversampling ratio (i.e., the ratio of the modulator clock to the Nyquist rate) low, which is desirable under certain conditions (col. 1, lines 25-35, see Cabler).
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the fourth-order cascaded sigma-delta modulator taught by Cabler with the electronic device of Massie in view of Kates in view of Davenport such that to obtain wherein quantization noise is shifted away from a first frequency range and to a second frequency range higher than the first frequency range in order to improve the performance of the cascade of modulators, as suggested by Cabler in paragraph column 3, lines 61-62.
Regarding claim 13, Massie in view of Kates in view of Davenport in view of Cabler teaches the filter circuitry of claim 12. Massie in view of Kates in view of Davenport in view of Cabler, as modified, teaches wherein the delta sigma modulator or the sigma delta modulator comprises a modulation order greater than one (Depending upon the number of integration stages, sigma-delta modulators can be divided into order types, e.g., second-order, third-order, or fourth-order, col. 1, lines 15-18, see Cabler).
Regarding claim 14, Massie in view of Kates in view of Davenport in view of Cabler teaches the filter circuitry of claim 12. Massie in view of Kates in view of Davenport in view of Cabler, as modified, capable teaches wherein modulating the intermediate signal comprises change from the second bit-depth to a third-bit depth, of the filter input signal, less than the second bit-depth by increasing the sampling rate (As explained earlier, for a fixed power and cost, decreasing the sampling rate enables us to choose a higher bit-rate quantizer. To reflect this, we examine two cases defined by setting such that the Nyquist-rate sampler starts at: (i) b= 4bits and (ii) b = 8 bits. From this experiment we see that in both cases, the RSNR grows significantly, achieving a 20 dB gain at 4 octaves of subsampling over Nyquist sampling in the 4 bit case and a 17 dB gain in the 8 bit case. The performance then decreases as we move to a regime where CS recovery is no longer sustainable. The oracle performance continues to improve as subsampling is further increased. This experiment highlights the very real benefit of reduced sampling rates; easing the sampling rate requirement can allow us to use higher fidelity hardware components, such as high bit-depth quantizers (page 4638, right column, second and third paragraphs, see Davenport). The motivation is in order to provide a straightforward tradeoff between noise folding (bad) and dynamic range in CS system, as suggested by Davenport in page 4638, right column, last paragraph.
16. Claim 15 is rejected under 35 U.S.C. 103 as being unpatentable over Massie et al. U.S. Patent 8611551 (hereinafter, “Massie”) in view of Kates U.S. Patent Application Publication 20060008101 in view of Davenport et al. “The Pros and Cons of Compressive Sensing for Wideband Signal Acquisition: Noise Folding versus Dynamic Range”, IEEE Transactions on Signal Processing, Vol. 60, No. 9, September 2012, 4628-4642, 15 pages (hereinafter, “Davenport”), and further in view of Park et al. Application Publication 20170229110 (hereinafter, “Park”).
Regarding claim 15, Massie in view of Kates in view of Davenport teaches the electronic device of claim 11. Massie in view of Kates in view of Davenport, as modified, teaches wherein the input signal is indicative of an environment audio sound (FIG. 1 is an illustration of an environment in which embodiments of the present technology may be used. An audio device 104 may act as a source of audio content to a headset 120 which can be worn over or in the ears 103, 105 of a user 102, Fig. 1, col. 3, line 64 – col. 4, line 1, see Massie) and the filtered signal is indicative of an inverted audio sound (In order to achieve a relatively low latency, an ANC system may be implemented using analog filter circuitry. The analog circuitry filters and inverts the analog reference signal received from the reference microphone to form an analog compensation signal, which is then provided to the loudspeaker, col. 2, lines 3-8, see Massie), wherein the filter circuitry is configured to receive a feedback signal (In addition, it is possible to combine both microphone inputs m(t) and r(t) in a mixed feedforward/feedback technique that includes two filtering blocks (one that implements the feedforward part of the processing, and one that implements the feedback part of the processing, col. 10, lines 44-48, see Massie).
However, Massie in view of Kates in view of Davenport does not explicitly disclose wherein the filter circuitry is configured to: receive a feedback signal indicative of residual sound comprising a summation of the environment audio sound and the inverted audio sound; and adjust one or more coefficients of the set of filter coefficients based on the feedback signal.
Park teaches systems, methods, apparatus, and computer-readable media for adaptive active noise cancellation (see Title) in which active noise cancellation (ANC, also called active noise reduction) is a technology that actively reduces acoustic noise in the air by generating a waveform that is an inverse form of the noise wave (e.g., having the same level and an inverted phase), also called an “antiphase” or “anti-noise” waveform. An ANC system generally uses one or more microphones to pick up an external noise reference signal, generates an anti-noise waveform from the noise reference signal, and reproduces the anti-noise waveform through one or more loudspeakers. This anti-noise waveform interferes destructively with the original noise wave to reduce the level of the noise that reaches the ear of the user (par [0005], see Park). FIG. 1B shows a block diagram of an example A20 of an ANC apparatus that includes a feedback ANC filter F20 and an error microphone ME10 that is disposed to sense sound at a user's ear canal, including sound (e.g., an acoustic signal based on antinoise signal SY10) produced by loudspeaker LS10. Filter F20 is arranged to receive an error signal SE10 that is based on a signal produced by error microphone ME10 and to produce a corresponding antinoise signal SY10 (Fig. 1B, par [0073], see Park). In an implementation of ANC filter F70 that includes an IIR filter, one or more (possibly all) of the feedforward filter coefficients and/or one or more (possibly all) of the feedback filter coefficients may be adaptable (i.e., adjust). Feedback ANC filter AF20 may be implemented as an adaptable filter according to the same principles discussed above with reference to FIG. 4C (Fig. 4C, par [0083], see Park).
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the systems, methods, apparatus, and computer-readable media for adaptive active noise cancellation taught by Park with the filter circuitry of Massie in view of Kates in view of Davenport such that to obtain wherein the filter circuitry is configured to: receive a feedback signal indicative of residual sound comprising a summation of the environment audio sound and the inverted audio sound; and adjust one or more coefficients of the set of filter coefficients based on the feedback signal for purpose of enhancing or separating clear desired sound from background sounds originating from multiple directions, as suggested by Park in paragraph [0182].
17. Claim 16 is rejected under 35 U.S.C. 103 as being unpatentable over Massie et al. U.S. Patent 8611551 (hereinafter, “Massie”) in view of Kates U.S. Patent Application Publication 20060008101 in view of Davenport et al. “The Pros and Cons of Compressive Sensing for Wideband Signal Acquisition: Noise Folding versus Dynamic Range”, IEEE Transactions on Signal Processing, Vol. 60, No. 9, September 2012, 4628-4642, 15 pages (hereinafter, “Davenport”), and further in view of Norgaard et al. U.S. Patent Application Publication 20200107138.
Regarding claim 16, Massie in view of Kates in view of Davenport teaches the electronic device of claim 11. However, Massie in view of Kates in view of Davenport does not explicitly disclose wherein the warped finite impulse response filter comprises a plurality of all-pass filter stages disposed in a transpose form.
Norgaard teaches method of estimating a feedback path of a hearing aid and a hearing aid (see Title) in which according to an alternative embodiment the feedback suppression filter 204 is a warped FIR filter, i.e. a filter with a frequency dependent delay and thereby a non-uniform frequency resolution as opposed to the traditional FIR filter that provides a uniform frequency resolution (Fig. 4, par [0052], see Norgaard). Following the same procedure as outlined above for the FIR filter implementation we find that an estimate custom-character of the warped filter coefficient vector may be determined (see superscript T in equation in par [0055], of Norgaard); where yT is the transposed input signal vector (see claim 6 of Norgaard on page 5, left column).
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the method of estimating a feedback path of a hearing aid and a hearing aid taught by Norgaard with the electronic device of Massie in view of Kates in view of Davenport such that to obtain wherein the warped finite impulse response filter comprises a plurality of all-pass filter stages disposed in a transpose form in order to provide an improved method of estimating a feedback path of a hearing aid with respect to especially speed, as suggested by Norgaard in paragraph [002].
18. Claim 20 is rejected under 35 U.S.C. 103 as being unpatentable over Massie et al. U.S. Patent 8611551 (hereinafter, “Massie”) in view of Kates U.S. Patent Application Publication 20060008101 in view of Davenport et al. "The Pros and Cons of Compressive Sensing for Wideband Signal Acquisition: Noise Folding versus Dynamic Range", IEEE Transactions on Signal Processing, Vol. 60, No. 9, September 2012, 4628-4642, 15 pages (hereinafter, “Davenport”), and further in view of Oliver U.S. Patent Application Publication 20100266143.
Regarding claim 20, Massie in view of Kates in view of Davenport teaches the non-transitory, machine-readable medium of claim 17. Massie in view of Kates in view of Davenport, as modified, teaches wherein filtering the filter input signal (the method comprising the steps of receiving and converting analog input signals to digital input signals, passing digital input signals through a plurality of cascaded all-pass filters, par [0009]; see also Fig. 8, see Kates) comprises: multiplying the filter input signal by a multiplier coefficient of the set of filter coefficients to generate a first transitional signal of a first filter stage of the warped finite impulse response filter (Multiply Input and Each AP Output Sequence by the Corresponding Filter Coefficient and Sum to Give the Filtered Output Sequence (see Fig. 6, see Kates); In at least one embodiment of the invention, the processing system is a warped FIR side-branch system consisting of a 16-sample data buffer and a 32-point warped FFT used in conjunction with a 31-tap warped FIR filter. The input data segment, after passing through the all-pass stages that provide the frequency warping, is windowed with a 32-point hanning (von Hann) window and no frequency-domain smoothing is applied to the spectrum, par [0040], see Kates).
However, Massie in view of Kates in view of Davenport, does not explicitly disclose comprising: adding the first transitional signal to a first all-pass output of a second filter stage of the warped finite impulse response filter to generate a second transitional signal; and applying an all-pass filter of the first filter stage to the second transitional signal to generate a second all-pass output signal.
Oliver teaches frequency-warped audio equalizer (see Title) in which comprising: multiplying, via the multiplier of a first filter stage of the plurality of filter stages, the filter input by a respective coefficient of the multiplier of the first filter stage (Four multipliers 640, 642, 644, and 646 are shown in communication with the all-pass filters 601-611. These multipliers represent coefficients h.sub.0, h.sub.1, h.sub.2, and h.sub.3 of the equalization filter 600b, Fig. 6B, par [0070], see Oliver) to generate a first transitional signal (Each filter 601-611 in the example equalization filter 600b includes the same warping factor .rho., as represented by multipliers 448, 504, 514, 604, 612, and 620 (corresponds to a first transitional signal), Fig. 6B, par [0073], see Oliver); adding the first transitional signal to a first all-pass output of the all-pass filter of a second filter stage of the plurality of filter stages to generate a second transitional signal (n more detail, the output 637 of the filter 607 is combined with the output 633 of the filter 603 by adder 630, which provides an output to the multiplier 642 (coefficient h.sub.1) (corresponds to a second transitional signal), Fig. 6B, par [0071], see Oliver); and applying the all-pass filter of the first filter stage to the second transitional signal to generate a second all-pass output signal (Likewise, the output 639 of the filter 609 is combined with the output 631 of the filter 601 and is provided to adder 632, which provides an output to the multiplier 644 (coefficient h.sub.2), Fig. 6B, par [0071], see Oliver). Since “the outputs of the filters 601-611 are therefore superimposed together (see Fig. 6B, par [0071])” thus output of each adder 630, 632 being considered as “transitional signal”.
It would have been obvious to one of ordinary skill in the art before the effective filing date of the claimed invention to incorporate the frequency-warped audio equalizer taught by Oliver with the non-transitory, machine-readable medium of Massie in view of Kates in view of Davenport such that to obtain comprising: multiplying, via the multiplier of a first filter stage of the plurality of filter stages, the filter input by a respective coefficient of the multiplier of the first filter stage to generate a first transitional signal; adding the first transitional signal to a first all-pass output of the all-pass filter of a second filter stage of the plurality of filter stages to generate a second transitional signal; and applying the all-pass filter of the first filter stage to the second transitional signal to generate a second all-pass output signal in order to accomplish the same task with improved accuracy on the same processor, as suggested by Oliver in paragraph [0086].
Conclusion
19. The prior art made of record and not relied upon is considered pertinent to applicant's disclosure.
Inventor
Publication
Number
Disclosure
Trotter et al
US Patent Application Publication
20050093727
Additionally, delta-sigma modulators perform noise shaping by acting as a high-pass filter to the noise such that most of the quantization noise power is shifted out of the signal band of interest (paragraph [0009]).
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/C.P.T/Examiner, Art Unit 2695
/VIVIAN C CHIN/Supervisory Patent Examiner, Art Unit 2695
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